Integrated Magnetic Components for Current Doubler Rectifier and Analogous Circuits Using Bobbin less U Cores

　　H. Njiende, R. Robrecht, P. Ide

　　Delta Energy Systems GmbH, Coesterweg 45, 59494 Soest, Germany

　　Tel.: +49 2921 987 472 Fax.: +49 2921 987 416

　　Email: hugues.njiende@delta-es.com

　　Abstract

　　Switched mode power supplies as main part of telecom and commercial systems often dictate their size and electrical performance as well as reliability and costs. As requirements for the key characteristics power density and efficiency of power converters increase, the demands of these evaluation characteristics increase for inductive components particularly. One powerful approach of increasing the power density and the efficiency is to integrate the inductive components. Transformers and inductors can be integrated into a single magnetic structure which than reduces cost, increases power density and power efficiency. Several published works on current- doubler rectifiers with integrated magnetic components are known, which are all based on EE cores with bobbin [1], [2], [3],[4], [5] and [6]. One additional approach of increasing the power density and the efficiency is the use of bobbin less cores. This paper presents new integrated magnetic structure composed of bobbin less U cores which are used as flexible building blocks. The core structure can either be fully composed of high permeability low saturation cores with air gaps or be composite comprising low permeability high saturation cores and high permeability low saturation cores with no air gaps. The operation and performance of the magnetic structure is verified on a 1KW, 400 V input 12V output Full-bridge converter.

　　1. Introduction

　　The current- doubler rectifier circuit, habitually applied for low voltage and high current outputs, is used with different double-ended primary topologies such as forward, two transistors- forward, push-pull, half bridge or full bridge inverters and uses one simple two- winding transformer and two output inductors (Fig.1 a). The current- doubler rectifier then exhibits lower conduction losses than the conventional center tapped rectifier. This configuration results, additionally to the number of discrete magnetic components which yield higher size and costs, in three high current windings and several high interconnection losses which negatively impact the efficiency. Consequently integrated magnetics are strongly recommended for this circuit to solve the listed problems. Several published works on current- doubler rectifiers with integrated magnetic components are known which are all based on EE cores with bobbin [1], [2] and [3].Resulting structure from [3] is depicted in Fig.2 b. Although these core forms are very popular and widely-used they lack flexibility for integrated magnetics. Moreover the bobbins lead to more leakage inductances and copper losses.

　　In the present work new bobbin less U cores are used on whose legs the windings are directly wound.The tight core- winding coupling yields lower leakage, minimized copper power losses and inductance losses as well as minimized overall thermal resistance. The U cores are used as building blocks which make their mechanical and magnetic (for ex. different materials) assembly more flexible. The core structure can either be fully composed of high permeability low saturation cores with air gaps or be composite comprising low permeability cores and high permeability cores with no air gaps.

　　A synthesis of the integration of cores and windings and the analysis of each integration step is performed. Beginning with discrete transformer and output filter inductors, the later are coupled properly to provide inverse coupling Fig. 2 a. Further the transformer and the coupled output inductors are integrated into a single structure, Fig.4 a. The transformer cores and secondary windings are removed without substantially altering the converter functionality, Fig.4 a and b. In order to achieve a reasonable value (1.35uH) of filter inductance without saturation of inner or outer core legs, additional filter windings (1 turn) are inserted and wound in parallel around outer core legs as illustrated in Fig.4 c.

　　The operation and performance of the final magnetic structure as well as its analysis is verified on a 1KW, 400 V input 12V output Full-bridge converter. Experiments results are presented in section 3.

　　2. From discrete magnetics to new integrated magnetics

　　The current- doubler rectifier, habitually applied for low voltage and high current outputs, is used with a full bridge inverter and comprises one simple two- winding transformer and two output inductors, (Fig.1 a). This converter circuit results, additionally to the number of discrete magnetic components, which yield higher size and costs, in three high current windings and several high interconnection losses which negatively impact the efficiency. The integrated magnetics technique constitutes an important tool in order to reduce the number of components and therefore the number of interconnections which results in lower winding and interconnection losses as well as lower material costs. Additionally the bobbin less technique is applied to furthermore reduce the winding losses through shortened mean turn length of winding and reduce leakage inductances through a tight coupling between windings and cores. Both techniques are combined to a powerful instrument for building of flexible, high density, high efficiency, and low cost magnetic component assembly.

　　Using the bobbin less U cores the discrete magnetic components are built. The primary and secondary windings of the transformer are split and each half winding wound around a core leg tightly (P1 and S1 on U1 as well as P2 and S2 on U2) in order to reduce proximity losses. The increase of coupled capacitance between windings and core is the only setback resulting from the tight coupling. For purpose of illustration the inductor windings are wound around just one core leg (L1 on U5 as well as L2 and U6) which is jointed to another core through distributed air gaps (g11 and g12 or g21 and g12). The distributed air gaps lead to reduction of arc of fringing fields which therefore result to a decrease of negative effects on nearby windings and components. In addition, contrarily to widely used EE cores, the air gaps are not perpendicular to the windings, which results in reduced fringing field losses [8]. Actually the air gap fringing field slightly affect just the end turns of the windings.

　　Adding to cited setbacks, the number of components and of high current interconnections, the current-doubler rectifier exhibits additional limitations. In order to reduce output ripple voltage the ripple currents of the inductors are reduced by increasing their inductances which implies more copper losses. On the other hand in order to allow a fast transient response during change in power requirements, the inductances need to be small. High efficiency and fast transient response requirements are conflicting regarding the design of the inductors [4], [5], [6] and [7]. Coupled inductors are suitable to achieve both requirements simultaneously.

　　2.1. Coupled output inductors

　　The first integration step is the coupling of both output inductors whereby the coupling is either direct or inverse [5], depending on the winding direction of inductor windings L1 und L2, Fig. 2 a

　　Applying Kirchhoff’ voltage Law on the reluctance model in Fig. 2 b, next matrix equation is defined R ⋅ Φ = N ⋅ I

　　with R and N .being reluctance and turn number matrices respectively. Vector of mesh fluxes and winding currents are as winding currents are as

　　Fig. 3. a) Winding current waveforms and b) core flux waveforms of the proposed structure in Fig. 2 a) Calculated ripples in Table 1 are used to plot current and flux waveforms, illustrated in Fig. 3 a and b, which strongly depend on the coupling coefficients k and QR, respectively. For uncoupled windings air gap in flanges between wounded core legs is large therefore QR→ 1 and k → 0 prevail. iL1 and iL2 are 180° phase shifted. Minimum current and flux ripples are achieved for minimum flange air gaps which means RL>> RF and RL>> RT and therefore QR→ 0 and k → 1 . For theoretically zero air gap between wounded core legs iL1 and iL2 are in phase as well as ÖL1 and ÖL2.

2.2.    Integrated transformer and output inductors

Though  the  windings  have  been  coupled  the  number  of  core  is  still  too  high, 6  U  cores. In  other  to reduce the  volume, transformer and inductor cores are integrated in the next step. Transformer primary and secondary windings are split and each primary and secondary half winding are tightly wound around a core with either of inductor windings as depicted in Fig. 4 a.  P₁,   S₁, and  L₁ are wound tightly around one core leg and  P₂,     S₂, and  L₂ as well around the other core leg. Current waveforms of primary and secondary transformer are similar and this structure still operates as current doubler rectifier with coupled inductors. This structure simultaneously transforms and stores energy: Energy is partly transformed in the  cores U₁ and  U₂ while partly stored in gapped cores U₃ andU₄.

he secondary winding is obsolete and can be removed without drastically altering the functionally of the integrated structure. This action yields the next integrated structure depicted in Fig. 4 b. Additionally to preceding core integration a winding integration is achieved. Now, contrarily to previous structures, the secondary  windings   S₁  and S₂  transform  and  store  energy. Either, current and  flux  waveforms  are different to those of previous structures: when diode  D₁ conducts and diode  D₂ blocks, the secondary winding  S₁ stores energy and secondary winding  S₂ is without current. The current and flux waveforms,shown  in  Fig. 5  a  and  b, are  now  similar  to  those  of  center  tapped  rectifier. Still, as  with  the  previous

integrated structure, cores U₁ and U₂ transform energy and gapped cores U₃ andU₄ store energy. For  low  voltage  and  high  current outputs, for  which  the  current doubler  rectifier  is  optimal, it is necessary  for  lowering  copper  losses  at full  load  to  have  lowest turn  number  at secondary  windings. Since  the  secondary  turn  number  also  account for  the  filter  inductor,  the  filtering  inductance  is  not  sufficient to  achieved  low  current ripples. A  remedy  to  this  setback  is  utilization  of  additional  filtering inductance  as  in  [3].  Additional  filter  winding  are  wound  around  outer  core  leg  as  depicted  in  Fig.  4  c.Output current is  distributed  between  paralleled  additional  windings  L_a1   and   L_a2 . Compared  to  the structure in [Sun], the current distribution generates temperature and manufacturing advantages: the heat dissipation is better and same conductor is used for all four secondary windings  S₁, S₂, L_a1  and  L_a2^.

　　In summary the additional windings cause in one hand a drastic increase of the resulting output filtering inductance L_fa  which  implies  reduced  ripple  output  current and  minimized  conduction  and  switching  losses. On  the  other  hand  these  windings  enable  a  reduction  of  ripple  fluxes  which  signifies  minimized cores losses. Although the air gaps are distributed, their inherited fringing fields still cause winding power losses and inductance  losses. Moreover, to  assure  mechanical  stability  a  non  permeable  material  is  inserted  at location of air gap. The mounting of this material negatively affects the bonding or gluing of the cores as well as the costs. As a solution to these hindrances, low permeability high saturation bobbins less U cores are proposed to replace the outer cores, U₃ and U₄, of the proposed integrated magnetic structures.In the following the integrated structure in Fig. 4 c is analyzed closely including description of functionality, calculation of different current and flux ripples as well as of current and flux waveforms. With reluctance and turn number matrices respectively as