Haaf, Peter, Fairchild Semiconductor, Germany, peter.haaf@fairchildsemi.com
Domeij, Martin, Fairchild Semiconductor, Sweden,martin.domeij@fairchildsemi.com
Abstract
Efficiency is becoming more and more important for the designers of power electronics today,
as well as size and cost. In boost DC/DC converters, typically used in PV inverters and
PFC circuits, increased switching frequency makes a big impact on both size and cost. Silicon
Carbide (SiC) bipolar junction transistors (BJT’s) offer low-loss high speed switching
combined with low conduction losses enabling higher switching frequency and maintaining
high efficiency. These devices turn off from the semi saturated state without current tails and
also have extremely low saturated Vce values. SiC BJT’s combine the best properties from
silicon unipolar and bipolar technologies in a normally-off device. The design and performance
of a 1kW boost circuit based on the SiC BJT is presented in this paper.
1. Synopsis: The SiC BJT
Silicon carbide (SiC) bipolar junction transistors (BJT) using a vertical NPN structure were
fabricated and assembled in an industrial standard TO-247 package. This type of transistor
combines very low conduction losses with fast and low loss switching behavior [1]. The high
critical field strength of silicon carbide gives the possibility to have low saturation voltages
without driving the transistor into hard saturation. Since there is no channel region in a BJT
the V is determined mainly by the collector series resistance.
It is not necessary to drive the SiC
BJT into hard saturation which
gives them excellent switching
properties. There is no current tailing
and a minimal storage delay (5
ns) at turn-off. The SiC BJT is also
a very robust device with a wide
RBSOA, good short circuit capability
and a minimal storage delay (5
ns) at turn-off. The SiC BJT is also
a very robust device with a wide
RBSOA, good short circuit capability
over time [4]. The BJT’s
discussed in this paper do not
have this problem (Figure 1).
2. DRIVE SOLUTIONS
2.1. Base Drive Circuit and the Challenges
A SiC bipolar transistor is a current driven device. The driver must deliver enough current to
turn the device on and off, plus load and unload the Miller charges quickly enough. Nevertheless
the losses in the drive circuit should be limited.
One target for this development was to decrease the driver losses as much as possible,
without impacting the switching speed.
To have enough safety margin the base current should be oversized by a factor of 1.5 compared
with the calculated base current, Ib = beta * Ic. To simplify the base drive circuit using
a constant base current is recommended.
We developed a driver board with an adjustable dual supply, Vcc and Vee, a high speed opto
isolator and an additional capacitor C10 in parallel to the base resistor R2 to boost the
base current for a short moment, just during turn-on and turn-off of the BJT.
Fig. 2 Base drive circuit for SiC BJT
When the BJT is turned on, Vbe is equal to 2.9V with a slightly negative temperature coeffici
ent.The positive supply voltage of the driver is divided into: a small voltage drop in the
driver, the Vbe and the voltage drop across the base resistance R2. The capacitor C10 in
parallel to R2 will be charged in this moment to the voltage across R2. When the device is
turned off, this capacitance provides a short increase in the drive current to increase the
switching speed of the BJT.
Two big obstacles for fast switching are the parasitic inductances of the package and the
PCB. To reduce the influence of these parasitics the supply voltage for the driver could be
increased, but this would lead to significant higher drive losses during turn-on.
The alternative would be two separate drivers, one for fast switching, the other for supplying
the base current. As this would make the circuit expensive and complicated, we decided to
use the additional boost capacitor C10 in parallel to the base resistance R2.
In any case, the negative supply for the driver cannot be avoided for fast switching.
2.2. Double Pulse Test Setup
The base drive circuit has been optimized with the double pulse test setup. Turn-on and turnoff
waveforms can be analyzed separately without the need for a full thermal design.
The test set up is optimized: the inductor is a 1mH coreless choke and the capacitors have
been placed in shortest distance to the semiconductors in order to reduce the current loop
inductance.
By changing the position of the diode, the same test setup can be used for the boost function.
In this case the diode will charge an additional bank of capacitors connected to the
load. That means that at least the primary side circuit and the drive circuit are identical and
the test results can be compared easily.
Fig. 3 Double pulse test setup
3. Double Pulse Test Results
Some differences from SiC BJTs to conventional switches are characterized by the turn-on
and turn-off waveforms shown in Figure 4 and 5.
The drive circuit is supplied by Vcc = 7V and Vee = -3V. The red curve M1 shows Vbe, which
is measured by the differential signal between base and emitter lead.
When turning the BJT on, the signal starts to rise from -3V to +3V by charging Cbe. At the
threshold of 3V the current starts to commutate from the diode to the BJT at a di/dt of
around 1000A/us.
The drive circuit is supplied by Vcc = 7V and Vee = -3V. The red curve M1 shows Vbe, which
is measured by the differential signal between base and emitter lead.
When turning the BJT on, the signal starts to rise from -3V to +3V by charging Cbe. At the
threshold of 3V the current starts to commutate from the diode to the BJT at a di/dt of
around 1000A/us.
The first drop of Vce is caused by the induced voltage across the loop inductor:
Vind = -Lloop*di/dt
As soon as the diode current drops to zero, the Vce of the BJT starts to drop. A strong base
driver is now needed to discharge the Miller capacitance of the BJT.
During turn-off, both the current and voltage waveform start to change at the same time. In
the first nanoseconds the base current must charge the Miller capacitance. As soon as Vce
reaches the bus voltage, the change in the di/dt of the current drop is visible.
Due to the ideal features of the SiC material the influence of the removal of the stored carrier
charge is negligible, this causes just a delay of about 5ns in the begin of the turn-off procedure.
3.1. Influence of parallel capacitors
Some capacitors are used in snubber circuits in order to solve EMI issues. In other cases,
like in inductive heating applications capacitors in parallel to the switch reduce the total
switching losses. Small parasitic capacitors can never be avoided. In conclusion it is important
to know the behavior of a fast switching device with paralleled capacitors.
We review the influence of a 100pF and a 470pF capacitance in parallel to the switch, which
has been added to the double pulse tester.
The Eon losses dominate the total switching losses. It is obvious that the reduction of the
Eoff losses cannot compensate for the increase of the Eon losses.
Only ZVS applications, like LLC and phase shift power supplies and BCM PFC circuits can
profit from similar conditions.
3.2. Influence of the package
It is often said that TO247 packages cannot handle the high switching speeds of the SiC devi
ces.
In order to double check we performed some tests as follows. As the driver board is
opto-isolated, the connection to the BJT is rather flexible. The GND-trace of the driver is
connected to two different points of the emitter lead of the BJT (Figure 8, option1 and 2); in
figure 9 the voltage drop across the emitter lead is measured.
The most significant difference is visible at highest currents and at maximum di/dt. During
turn-on at a di/dt of 1000A/us, we measured an increase of Eon losses from 117uJ to 172uJ
and the peak voltage drop across this lead has been measured at 7Vpeak. A similar effect
can presumably be seen when considering the voltage drop across the emitter bond wire
under the similar conditions. The good thing is the relatively large base-emitter capacitance
which prevents the device from a parasitic turn-off.
As a conclusion, the usage of modules with separated power-emitter pins and drive-emitter
pin will solve this problem and will make the device easier to control.
4. The Boost Converter
Three different boost configurations were reviewed. The first circuit was a continuous conduction
mode (CCM) boost circuit using with the 12A/1200V SiC BJT and 20A/1200V SiC diode.
This was tested at two switching frequencies: 20 kHz and 40 kHz.
The second circuit was a boundary conduction mode (BCM) boost circuit using the same devices.
The ZVS approach of the BCM boost looks very promising in comparison to the hard
switched CCM mode.
Finally we operated our high speed IGBT FGL40N120AND in combination with the SiC diode
at 20 kHz in CCM mode as a reference.
4.1. Boost in CCM
For the CCM operation we used a 10mH coreless inductor.
In order to reduce the drive losses the positive operation voltage the driver Vcc was reduced
to 6V. The negative driver supply Vee has been increased to Vee = -6V without any impact
on the drive losses.
The waveforms at max power at Vout = 812V are shown in Fig. 10 and 11
4.2. Boost in BCM
The big obstacle for BCM operation is high ripple currents and varying frequencies. A typical
power range of a practical 2-phase interleaved PFC application is about 1kW down to
90VAC input voltage. Increasing the input voltage to 400VAC, it is easily possible to increase
the power range to 4kW without changing the current level. By going from PFC to boost or
from 2-phase to 4-phase interleaved operation a power level of 8kW with this topology will
be no problem.
Figures 12 and 13 show the test results.
The very smooth decrease of the Vce and the turn-on without any losses is clearly visible.
The parasitic capacitance of the inductor and the junction capacitance of the SiC diode limit
the dV/dt in Fig. 13 to 5kV/us. All transitions look very smooth and promising, especially regarding
EMI performance. For the inductor we chose a Kaschke E-core with
L= 0.68uH and multistrand wire, which is optimized for high di/dt.
4.3. Reference boost with IGBT in CCM
Going back to CCM mode, we would like to compare the very well known behavior of a fast
switching 1200V IGBT to the SiC BJT. The identical power circuit operates at 20kHz, the
driver is as well the FAN3122, but for IGBTs it operates between Vcc = 15V and GND. The
gate resistor Rg is equal to 4.7Ohm. The IGBT at a TB=100ns shows about 10x higher Eoff
losses (Figure 14). The Eon losses of the SiC BJT and IGBT are in the same range.
4.4. Efficiency Results
The boost measurements has been performed with a constant Vin = 210VDC. The output
voltage was adjusted by changing the duty cycle or/and the frequency in 100V steps from
Vout = 400V to Vout = 800V. The load is a parallel and series operation of incandescent
bulbs, total: 880W
Fig. 16 measured efficiency of the variations of the boost converters
The BCM boost shows best results, the CCM boost at 20kHz and 40kHz medium and the SiIGBT
comes third in this competition. The higher efficiency for the 40kHz CCM circuit versus
the 20kHz CCM is attributed to the lower ripple current seen in the inductor and capacitor at
40kHz compared with the 20kHz solution.
5. Analysis
Actually we would have expected in general a better total efficiency for all measurements.
Similar to tests seen on very low RDS(ON)MOSFETs in DC/DC applications, the package and
layout effects start to dominate as the losses of the semiconductors become less significant.
Based on the accompanying scope measurements for Eon and Eoff, the loss calculation for
the switches has been performed. Based on datasheets and temperature the assumption for
conduction losses was: SIC BJT, Ron = 100mOhm; IGBT, VCE(SAT) = 1V.
Table 1 Loss calculation of the switches in the application at Vin=210V, Vout=800V, Pout=735W
The weak point of the test setup is a symmetric EMI filter, which should prevent noise getting
in the power analyser. A big portion of the total losses are dissipated in this device. More
than 80% of the losses are dissipated in the passive components.
6. Conclusion
The SiC BJT offers significant efficiency advantages in boost applications compared with silicon
IGBT technology. Similar to when using other high performance technologies, the passive
and layout parasitic elements start to dominate the losses.
7. Literature
[1] M. Domeij et. al., “2.2 kV SiC BJTs with low VCESAT and fast switching” to be published
in the Proceedings of the International Conference on Silicon Carbide and Related Materials
(ICSCRM),2009.
[2] Y. Gao et. al., “Comparison of static and switching characteristics of 1200 V 4H-SiC BJT
and 1200 V Si-IGBT“, IEEE Transactions on Industry Applications, Volume 44, No. 3, p.
887, 2008
[3] A. Lindgren, M. Domeij, “1200 V 6 A High Temperature SiC BJTs”, Proceedings of HiTEC
Conference, 2009.
[4] M. Domeij et. al., “Degradation-free 1200 V 6 A SiC BJTs with very low VCESAT and
fast switching”, Proceedings of PCIM Conference, 2009.
[5] A. Lindgren, M. Domeij, “Degradation free fast switching 1200 V 50 A Silicon Carbide
APEC, 2009.
